Semiconductor integrated circuit

ABSTRACT

During a period of preparation for actual operation, a reference clock is supplied to both a comparison clock input portion and a feedback clock input portion of a phase comparator while a feedback loop of a PLL (phase-locked loop) is interrupted, and a delay of a reset signal within the phase comparator is adjusted so as to reduce a detection dead zone of phase differences in the phase comparator.

BACKGROUND OF THE INVENTION

The present invention relates to semiconductor integrated circuitsincluding a clock generation circuit, and more particularly tosemiconductor integrated circuits in which a PLL (phase-locked loop)circuit is provided.

Computing systems, such as microprocessors or microcontrollers, areprovided with PLL circuits functioning as a clock multiplier circuit inorder to realize the function of multiplying an external frequency at aportion of the central processing unit to perform high-speed operations.Furthermore, in recent microprocessors, it is desirable to be able tomaintain the clock phase between an external bus and within thesemiconductor integrated circuit with high precision.

In conventional methods, the time that it takes until the PLL circuitstabilizes after the power has been turned on is counted with a timer,the clock supply from the PLL circuit to the central processing unit ishalted for a certain amount of time, and the multiplied clock supply isstarted as soon as the timer overflows.

Now, in the phase comparator of the PLL circuit it is desirable thatthere is a linear relation between the phase difference of the twosignals entered into it and the voltage that is output. However inpractice, there are cases in which it is not possible to detect tinyphase differences, so that there may be a dead zone of phasedifferences, and there may be discontinuities when the sensitivity istoo high.

It is known that the length of the delay time in the reset circuit has alarge influence on the input/output characteristics of the phasecomparator. In other words, in order to improve the input/outputcharacteristics of the phase comparator, it is necessary to adjust thedelay time in the reset circuit to an appropriate value. However, in thephase comparator according to a first piece of conventional technology,the delay time becomes shorter than the appropriate value because thereset circuit is made of one 4-input NAND circuit, and the input/outputcharacteristics exhibit a dead zone (U.S. Pat. No. 3,610,954).

Several improvements have been suggested in order to adjust the delaytime of the reset circuit to an appropriate value. In a second piece ofconventional technology, the output of the reset signal is delayed bymaking the channel width of a transistor constituting the 4-input NANDcircuit narrower (JP S63-119318A). Furthermore, in a third piece ofconventional technology, a plurality of capacitors are used as a meansfor delaying the output of the reset signal (U.S. Pat. No. 4,378,509).

As described above, in the phase comparator according to the firstconventional technology, the reset circuit is constituted by one 4-inputNAND circuit, so that the delay time becomes shorter than theappropriate value and there is a dead zone in the input/outputcharacteristics. In the case of the second conventional technology, aworsening of the yield due to variations in the channel width or thelike has become unavoidable with the sub-micron gate widths of recenttransistors. And with the third conventional technology, the capacitorslead to an increase of the chip surface area.

Charge pump circuits also have an aspect that worsens their input/outputcharacteristics. When using a current-type charge pump circuit, itoccurs that the output voltage of the phase comparator changes eventhough there is no phase difference between the two input signals. Thismeans that even though clocks of the same phase are input, the phasedifference is detected erroneously and a highly accurate PLL circuitcannot be realized.

Furthermore, clock drivers are designed such that they can supply aclock synchronized with zero skew to the function blocks, but due totemperature dependencies, process variations and the like, there areskew variations among chips.

Also, inside the function blocks, circuits that use two phases of clockswith clock synchronization, such as dynamic circuits or memories, aredesigned such that they can operate stably with some delay so as toavoid signal racing, but due to process variations, the margin betweenthe two phases of the clocks may disappear, resulting in faultyoperation.

Furthermore, there are function blocks that include the function ofinterrupting a series of operations when processing has becomeunnecessary during that series of operations, in order to reduce energyconsumption, but depending on the operation frequency and processvariations, the operation may not be halted completely, resulting infaulty operation.

Moreover, providing a tuning circuit in order to solve these problems isa waste of time, because the start of the operation of the tuningcircuit needs to wait until the PLL circuit has stabilized.

SUMMARY OF THE INVENTION

It is an object of the present invention to make it possible toefficiently utilize the time before a clock generation circuit suppliesa system clock signal, and in particular the time until the oscillationof a PLL circuit has stabilized.

In order to achieve this object, a semiconductor integrated circuit inaccordance with the present invention includes a clock generationcircuit that generates a system clock signal from a reference clocksignal, and a specific circuit portion within the semiconductorintegrated circuit is adjusted using the reference clock signal beforethe clock generation circuit supplies the system clock signal. Inparticular in a semiconductor integrated circuit provided with a PLLcircuit, the specific circuit portion is adjusted using the referenceclock signal before the PLL circuit has reached stable oscillation.

More specifically, the reference clock signal is supplied to both acomparison clock input portion and a feedback clock input portion of thephase comparator while a feedback loop of the PLL circuit isinterrupted, and the delay of a reset signal within the phase comparatoris adjusted so as to reduce a detection dead zone of phase differencesin the phase comparator.

In the case of a bandgap reference circuit for supplying a referencevoltage to a current charge pump circuit within the PLL circuit, thereference clock signal is supplied to either a comparison clock inputportion or a feedback clock input portion of a phase comparator withinthe PLL circuit while a feedback loop of the PLL circuit is interrupted,and a phase correction amount of that bandgap reference circuit isadjusted such that the bandgap reference circuit does not oscillate.

In the case of a current charge pump circuit within the PLL circuit, thereference clock signal is supplied to either a comparison clock inputportion or a feedback clock input portion of a phase comparator withinthe PLL circuit while a feedback loop of the PLL circuit is interrupted,and the current driving ability of the current charge pump circuit isadjusted.

In the case of a clock distribution circuit for distributing the systemclock signal to a plurality of function blocks, skew between a pluralityof clock drivers within the clock distribution circuit is adjusted suchthat output clock skew of the clock distribution circuit is eliminated.

In the case of a data holding portion operating in synchronization withthe system clock signal, such as a memory circuit including a word lineand a sense amplifier, or a dynamic circuit of at least two stagesconnected in series, a racing adjustment is performed in the internaloperation of that data holding circuit.

In the case of a functional circuit having a power consumption reductionfunction such as a cache circuit, when it has been detected from thereference clock signal and a feedback clock signal of the PLL circuitthat a phase fine-tuning period has been entered after frequencycapturing of the PLL circuit has been terminated, an adjustment is madeby stopping the operation of one of the circuit portions within thefunctional circuit in correspondence with the oscillation clock signalof the PLL circuit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a semiconductor integrated circuit inaccordance with a first embodiment of the present invention.

FIG. 2 is a circuit diagram showing the configuration of the phasecomparator in FIG. 1.

FIG. 3 is a circuit diagram showing the configuration of the resetcontrol voltage circuit in FIG. 1.

FIG. 4 is a timing chart illustrating the operation of the semiconductorintegrated circuit in FIG. 1.

FIG. 5 is a block diagram of a semiconductor integrated circuit inaccordance with a second embodiment of the present invention.

FIG. 6 is a circuit diagram showing the configuration of the PLL circuitin FIG. 5.

FIG. 7 is a circuit diagram showing the configuration of the referencevoltage circuit in FIG. 5.

FIG. 8 is a circuit diagram showing the configuration of the switchcircuits in FIG. 5.

FIG. 9 is a timing chart illustrating the operation of the semiconductorintegrated circuit in FIG. 5.

FIG. 10 is a block diagram of a semiconductor integrated circuit inaccordance with a third embodiment of the present invention.

FIG. 11 is a circuit diagram showing the configuration of switchcircuits in FIG. 10.

FIG. 12 is a circuit diagram showing the configuration of other switchcircuits in FIG. 10.

FIG. 13 is a timing chart illustrating the operation of thesemiconductor integrated circuit in FIG. 10.

FIG. 14 is a block diagram of a semiconductor integrated circuit inaccordance with a fourth embodiment of the present invention.

FIG. 15 is a circuit diagram showing the configuration of the phasecomparator in FIG. 14.

FIG. 16 is a circuit diagram showing the configuration of the switchcircuit in FIG. 14.

FIG. 17 is a block diagram showing the configuration of the registercontrol circuit in FIG. 14.

FIG. 18 is a circuit diagram showing the configuration of the rippledetection termination circuit in FIG. 17.

FIG. 19 is a timing chart illustrating the operation of thesemiconductor integrated circuit in FIG. 14.

FIG. 20 is a block diagram of a semiconductor integrated circuit inaccordance with a fifth embodiment of the present invention.

FIG. 21 is a block diagram showing the configuration of a memory accesscircuit in FIG. 20.

FIG. 22 is a circuit diagram showing the configuration of the dummy rowdecoder in FIG. 21.

FIG. 23 is a circuit diagram showing the configuration of the normal rowdecoders in FIG. 21.

FIG. 24 is a circuit diagram showing the configuration of the dummymemory cells in FIG. 21.

FIG. 25 is a circuit diagram showing the configuration of the normalmemory cells in FIG. 21.

FIG. 26 is a circuit diagram showing the unit configuration of the senseamplifying array in FIG. 20.

FIG. 27 is a circuit diagram showing the configuration of the comparatorin FIG. 20.

FIG. 28 is a block diagram showing the configuration of theincremental/decremental register in FIG. 20.

FIG. 29 is a timing chart illustrating the operation of thesemiconductor integrated circuit in FIG. 20.

FIG. 30 is a block diagram of a semiconductor integrated circuit inaccordance with a sixth embodiment of the present invention.

FIG. 31 is a timing chart illustrating the operation of thesemiconductor integrated circuit in FIG. 30.

FIG. 32 is a block diagram of a semiconductor integrated circuit inaccordance with a seventh embodiment of the present invention.

FIG. 33 is a circuit diagram showing the configuration of the phasefine-tuning period detection circuit in FIG. 32.

FIG. 34 is a circuit diagram showing the configuration of the switchcircuits in FIG. 32.

FIG. 35 is a timing chart illustrating the operation of thesemiconductor integrated circuit in FIG. 32.

DETAILED DESCRIPTION OF THE INVENTION

Referring to the accompanying drawings, the following is a detaileddescription of embodiments of a semiconductor integrated circuit inaccordance with the present invention.

Embodiment 1

FIG. 1 is an example of a semiconductor integrated circuit according tothe present invention, and is a block diagram illustrating aconfiguration example of a semiconductor integrated circuitincorporating a PLL circuit. In FIG. 1, numeral 50 denotes a PLL circuitincluding a phase comparator 51, a loop filter 52, a voltage-controlledoscillator 53, and a programmable frequency divider 54. The phasecomparator 51 has input ports Fp and Fr, and compares the phases of thesignals that are input into those two ports. A reference clock 100 isinput into Fp. The output 51 a of the phase comparator 51 is connectedto the loop filter 52, the output 52 a of the loop filter 52 isconnected to the voltage-controlled oscillator 53, and thevoltage-controlled oscillator 53 converts this input voltage into afrequency. The clock signal output from the voltage-controlledoscillator 53 is connected to the programmable frequency divider 54. Theswitch circuit 55 is controlled by a feedback control signal 3. Whenthis feedback control signal 3 is “H,” then Fr of the phase comparator51 is connected to the programmable frequency divider 54, and when thefeedback control signal 3 is “L,” then Fr of the phase comparator 51 isconnected to the reference clock signal Fp. In this example of theswitch circuit 55, numerals 6 and 7 denote N-type MOS (metal oxidesilicon) transistors, and numerals 5 and 8 denote P-type MOS (metaloxide silicon) transistors. The MOS transistors 5 and 6 togetherconstitute a transfer gate, as do the MOS transistors 7 and 8. Numeral 4denotes an inverter. The output 52 a of the loop filter 52 is given intoa reset control voltage generation circuit 1. This reset control voltagegeneration circuit 1 takes a PLL ON signal 56, which enables theoperation of the PLL circuit 50, as a reset signal, performs asynchronization operation using the reference clock 100, and detectsripples in the loop filter output 52 a. When there are ripples, itgenerates a voltage that is lower than the initial voltage, and outputsthis voltage as a reset control voltage, which is input into the phasecomparator 51. Moreover, if no ripples are detected, a voltage that ishigher than the initial voltage is generated as the reset controlvoltage 2.

FIG. 2 is an example of the phase comparator 51 of the PLL circuit 50,in which numeral 30 denotes a digital phase comparator, and numeral 40denotes a charge pump circuit. The digital phase comparator 30 is madeof a reset circuit 31, a first flip-flop 32, a second flip-flop 33, afirst 3-input NAND circuit 34, a second 3-input NAND circuit 35, a firstinverter 36, a first 2-input NAND circuit 37, a second inverter 38, anda second 2-input NAND circuit 39. The reference clock signal Fp is inputvia the first inverter 36 to the first NAND circuit 37, whereas thecomparison clock signal Fr is input into via the second inverter 38 intothe second NAND circuit 39. The output signal of the first NAND circuit37 is input into the first flip-flop 32 and the first 3-input NANDcircuit 34, whereas the output signal of the second NAND circuit 39 isinput into the second flip-flop 33 and the second 3-input NAND circuit35. The output signal of the first flip-flop 32 is input into the first3-input NAND circuit 34, whereas the output signal of the secondflip-flop 33 is input into the second 3-input NAND circuit 35. The resetcircuit 31 is made of a 4-input NAND circuit 31, into which are inputthe output signals of the first flip-flop 32 and the second flip-flop 33as well as the output signals of the first NAND circuit 37 and thesecond NAND circuit 39. The output signal of the reset circuit 31 isconnected to the source of a transfer gate 31 b, whose drain is input asa reset signal to the first flip-flop 32 and the second flip-flop 33,but is also input into the first 3-input NAND circuit 34 and the second3-input NAND circuit 35. The gate of the N-type MOS transistor of thetransfer gate 31 b is connected to the reset control voltage 2 inFIG. 1. The gate of the P-type MOS transistor of the transfer gate 31 bis connected to ground. When the potential of the reset control voltage2 becomes high, the output of the transfer gate 31 b changes faster, andwhen the potential of the reset control voltage 2 becomes low, theoutput of the transfer gate 31 b changes slower.

The first 3-input NAND circuit 34 outputs a first phase-differencedetection signal Pu, which is ordinarily “H,” but which becomes “L”while the phase of the reference clock signal Fp is ahead of thecomparison clock signal Fr. The second 3-input NAND circuit 35 outputs asecond phase-difference detection signal Pd, which is ordinarily “H,”but which becomes “L” while the phase of the reference clock signal lagsbehind the comparison clock signal Fr. The charge pump circuit 40 ismade of a P-type MOS transistor 41, an N-type MOS transistor 42 and aninverter 43. The source of the P-type MOS transistor 41 is connected toa current source, and its drain is connected to the drain of the N-typeMOS transistor 42. The source of the N-type MOS transistor 42 isconnected to ground. The first phase-difference detection signal Puoutput from the first 3-input NAND circuit 34 is input into the gate ofthe P-type MOS transistor 41, whereas the second phase-differencedetection signal Pd output from the second 3-input NAND circuit 35 isinput into the gate of the N-type MOS transistor 42, after beinginverted by the inverter 43. The drain of the P-type MOS transistor 41(and the drain of the N-type MOS transistor 42) is connected to theoutput terminal 51 a.

When the first phase-difference detection signal Pu is “L,” the P-typeMOS transistor 41 becomes conducting, so that the drain potential of theP-type MOS transistor 41 (potential of the output 51 a) increases. Andwhen the second phase-difference detection signal Pd is “L,” then theoutput signal of the inverter 43 becomes “H” and the N-type MOStransistor 42 becomes conducting, so that the drain potential of theP-type MOS transistor 42 (potential of the output 51 a) decreases. Thismeans that the potential of the output 51 a increases when the phase ofthe reference clock signal Fp is ahead of the comparison clock signalFr, and decreases when it lags.

FIG. 3 shows an example of the reset control voltage generation circuit1. The reset control voltage generation circuit 1 includes a rippledetection circuit 210, an incremental counter 230, an incrementalcounter 240, a ripple elimination termination circuit 220, and a resetcontrol voltage output circuit 250. The ripple detection circuit 210detects ripples in the ripple filter output 52 a. The incrementalcounter 230 is incremented when the ripple detection circuit 210 detectsripples. The incremental counter 240 is incremented when the rippledetection circuit 210 does not detect ripples. The ripple eliminationtermination circuit 220 sets the feedback control signal 3 to “H” andturns off the clocks of the ripple detection circuit 210 and theincremental counters 230 and 240 when, within three periods of thereference clock 100, ripples are detected only at the first and thethird period but not at the second period. When the incremental counter230 is incremented, the reset control voltage output circuit 250decreases the reset control voltage 2, and when the incremental counter240 is incremented, the reset control voltage output circuit 250increases the reset control voltage 2.

The ripple detection circuit 210 is made of P-type MOS transistors 211,212 and 213, an N-type MOS transistor 214, and a latch circuit 219 thatholds data during the period that the clock 218 is “L.” The rippledetection circuit 210 acts as a dynamic circuit, with a clock signal 229generated by the ripple elimination termination circuit 220. Thepotential of the voltage 216 is generated by the P-type MOS transistors211 and 212, at a desired voltage value. When the loop filter 52generates a voltage that is by the threshold of the N-type MOStransistor 214 higher than the potential of the voltage 216, then theoutput signal 215 of the ripple detection circuit 210 is changed from“H” to “L.” When no ripples are detected, the output signal 215 stays“H.”

The incremental counters 230 and 240 include half adders (HA) made ofEXOR circuits (exclusive or circuits: output is “H” only when input isinconsistent) 232, 236, 242, 245, and AND circuits 233, 237, 241, 244,as well as flip-flops 234, 235, 243, and 246 with reset. Numeral 259 inFIG. 3 denotes a 1-bit incremental register made of the low-order HA232, 232 and flip-flop 234 with reset, which receives the output 215 ofthe ripple detection circuit 210 via the inverter 231. The clock 218generated by the ripple elimination termination circuit 220 is inputinto the clocks of the flip-flops 234, 235, 243 and 246, and the PLLONsignal 56 is input into the resets of those flip-flops.

The ripple elimination termination circuit 220 is made of flip-flops 221and 222 with reset, EXOR circuits 223 and 227, a 3-input AND circuit224, an inverter 226, an AND circuit 225, and a buffer 228. The datainput into the flip-flop 221 is the output signal 215 of the rippledetection circuit 210, and the data input into the flip-flop 222 is theQ output of the flip-flop 221. The outputs of the flip-flops 221 and 222are input into the EXOR circuit 223, and the output of the flip-flop 221as well as the output signal 215 of the ripple detection circuit 210 areinput into the EXOR circuit 227. The output of the EXOR circuits 223 and227 and the output signal 215 of the ripple detection circuit 210 areinput into the 3-input AND circuit 224, and the output of the 3-inputAND circuit 224 is input into the inverter 226 and connected to thefeedback control signal 3. The output of the inverter 226 and thereference clock 100 are input into the AND circuit 225, and the outputof the AND circuit 225 is used as the clock 229, and connected to thebuffer 228. The output of the buffer 228 is used as the clock 218. Theclock 218 is used as the clock of the flip-flops 221 and 222, and thePLLON signal 56 is used as the reset of those flip-flops.

The reset control voltage output circuit 250 is made of a parallelarrangement of P-type MOS transistors 256, 255 and 254, and a parallelarrangement of N-type MOS transistors 251, 252 and 253. The gate lengthsof the P-type MOS transistors 256, 255 and 254 and the N-type MOStransistors 251, 252 and 253 are set to a ratio of 4:2:1. The gate of256 is connected to the output 238 of the flip-flop 234, and the gate of255 is connected to the output 239 of the flip-flop 235. The gate of 251is connected to the output of the flip-flop 243, which has been invertedinto the output signal 249 with an inverter 247. The gate of 252 isconnected to the output of the flip-flop 246, which has been invertedinto the output signal 257 with an inverter 248.

FIG. 4 is a timing chart of the signals in FIG. 1, FIG. 2 and FIG. 3. InFIG. 4, the horizontal axis denotes time, and the vertical axis denotesthe feedback control signal 3, the two input ports Fp and Fr of thephase comparator 51, the output 52 a of the loop filter 52, the rippledetection circuit output 215, the clock (clockb) 218, the 2-bit registerinternal states 221, 222, and, expressed in binary notation, theinternal states of the flip-flops 234 and 235 constituting theincremental counter 230 as well as the internal state of the flip-flops243 and 246 constituting the incremental counter 240, and the resetcontrol voltage 2.

Referring to FIG. 4, the following is an explanation of the operation ofFIG. 1, FIG. 2 and FIG. 3, constituting Embodiment 1. In the PLL circuit50, before the power is turned on, the PLLON signal 56 is “L,” and thevalue in the flip-flops 221, 222, 234, 235, 243 and 246 inside the resetcontrol voltage generation circuit 1 is “L.” After the power is turnedon, the PLLON signal 56 becomes “H,” and first, when the feedbackcontrol signal 3 is “L,” the feedback loop is interrupted, so that thereference clock 100 is input into Fr of the phase comparator 51 with thesame period and the same phase as into Fp. Ideally, if clocks of thesame phase are input into the phase comparator 51, no ripples shouldoccur in the loop filter output 52 a. However, in the case of thisexample, let us assume that due to process variations, the reset delaytime of the phase comparator 51 has become shorter than the desiredtime. Ripples occur in the loop filter output 52 a in the first periodof the reference clock 100. Thus, the output signal 215 of the rippledetection circuit 210 becomes “L,” so that “H” is input into thelow-order HA of the incremental counter 230, and the internal state ofthe flip-flops 234, 235 becomes “01”. Thus, the gate of the P-type MOStransistor 256 of the reset control voltage output circuit 250 becomes“H,” and the P-type MOS transistor 256 is cut off. Since the P-type MOStransistors 256, 255 and 254 are connected in parallel, their ONresistance becomes higher, and the potential of the reset controlvoltage decreases. This is transmitted to the gate electrode of thetransfer gate 31 b in FIG. 2, increasing its delay. As a result, in thesecond period of the reference clock 100, the delay of the reset outputof the digital phase comparator 30 is increased. In the second period,there are still ripples in the output of the loop filter 52, and thereset control voltage output circuit 250 further decreases the potentialof the reset control voltage 2. Thus, the delay of the reset output ofthe phase comparator 30 becomes even larger. In the third period, thereare no more ripples in the output of the loop filter 52. At the timewhen there are no more ripples, “H” is input into the incrementalregister 240 of the reset control voltage generation circuit 1. Then,the reset control voltage generation circuit 1 increases the potentialof the reset control voltage 2. In the fourth period, the delay of thereset output of the digital phase comparator 30 becomes smaller than inthe third period, and ripples start to appear again. The delay of thereset output of the digital phase comparator 30 becomes larger, and inthe fifth period, there are no more ripples in the output of the loopfilter 52. At the time when there are no more ripples, the output of theAND circuit 224 of the ripple elimination termination circuit 220 of thereset control voltage generation circuit 1, that is, the feedbackcontrol signal 3 becomes “H.” The internal clock 229 is stopped, and thepotential of the reset control voltage 2 is held. Then, in the sixthperiod, the PLL circuit 50 is connected by the switch circuit 55 to thefeedback loop, and ordinary stable PLL oscillation is reached. Thus, itis possible to realize a phase comparison that is very precise withrespect to temperature fluctuation and initial device variations of thedigital phase comparator 30.

It should be noted that in FIG. 2, due to variations in the switchingvoltage of the 3-input NAND circuits 34 and 35, there is the possibilitythat Pu and Pd are output at the same time, but it is also possible toease this by inserting a buffer between the transfer gate 31 b and the3-input NAND circuits 34 and 35 to make the output waveform steep.Furthermore, it is preferable that the delay time between the 3-inputNAND circuit 34 and the P-type MOS transistor 41 and the delay timebetween the 3-input NAND circuit 35 and the N-type MOS transistor 42 aremade the same by adjusting the transistor sizes or adding a buffer. Itis further possible to control not only the gate voltage of the N-typetransistor but also the gate voltage of the P-type transistor in thetransfer gate 31 b in FIG. 2.

The phase comparator 30 shown in FIG. 2 is only an example, and as longas it is a phase comparator that is configured by a sequential logicwith a reset function, the reset delay can be varied with a similarapproach with any type.

Embodiment 2

FIG. 5 is another example of a semiconductor integrated circuitaccording to the present invention. The semiconductor integrated circuitin FIG. 5 has a PLL circuit 500 and a reference voltage circuit 600. Theoutput of the charge pump circuit of the PLL circuit 500 is connected toa ripple detection circuit 900, and the output of the ripple detectioncircuit 900 is connected to a 2-bit incremental counter 910 that isincremented when ripples are detected. An output bus of this incrementalcounter 910 is connected to a control signal of switch circuits 930 thatrespectively connect En 626 to capacitors 920 and 921 when the controlinput e is “H,” and disconnect En 626 when the control input e is “L.”The capacitors 920 and 921 are respectively set to ¼ and ½ of thecapacitance of a capacitor 630 inside the reference voltage circuit 600.The ripple detection circuit 900 is the circuit 210 explained forEmbodiment 1, and also the incremental counter 910 is similar.

FIG. 6 is an example of the PLL circuit 500 according to the presentinvention. In FIG. 600, numeral 500 denotes a PLL circuit, which is madeof a phase comparator 51, a loop filter 52, a voltage-controlledoscillator 53, and a programmable frequency divider 54. The output ofthe phase comparator 51 is connected to the loop filter 52, and theoutput 52 a of the loop filter 52 is connected to the voltage-controlledoscillator. The voltage-controlled oscillator 53 converts its inputvoltage into a frequency. The clock signal that is output by thevoltage-controlled oscillator 53 is connected to the programmablefrequency divider 54. The switch circuit 55 is controlled by thefeedback control signal 3, and when the feedback control signal 3 is“H,” the Fr of the phase comparator 51 is connected to the programmablefrequency divider 54, whereas when the feedback control signal 3 is “L,”the Fr of the phase comparator 51 is connected to a switching circuit510. Using an input switch control signal 540, the switching circuit 510inputs the reference clock 100 into Fr of the phase comparator 51 onlywhen the input switch control signal 540 is “H,” and when it is “L,” itpegs the Fr of the phase comparator 51 to ground. In this example of theswitching circuit 510, numerals 515 and 518 denote N-type MOStransistors, and numerals 516 and 517 denote P-type MOS transistors. TheMOS transistors 515 and 516 together constitute a transfer gate, as dothe MOS transistors 517 and 518. Numeral 514 denotes an inverter. On theother hand, the reference clock Fp of the phase comparator 51 isconnected to the switching circuit 501. Using the input switch controlsignal 540, the switching circuit 501 inputs the reference clock 100into Fp of the phase comparator 51 only when the input switch controlsignal 540 is “L,” and when it is “H,” it pegs the Fr of the phasecomparator 51 to ground. In this example of the switching circuit 501,numerals 505 and 508 denote N-type MOS transistors, and numerals 506 and507 denote P-type MOS transistors. The MOS transistors 505 and 506together constitute a transfer gate, as do the MOS transistors 507 and508. Numeral 504 denotes an inverter. Furthermore, in FIG. 6, the phasecomparator 51 is separated into a digital phase comparator 30 and acurrent charge pump circuit 520. The current charge pump circuit 520 ismade of P-type MOS transistors 521 and 523, N-type MOS transistors 524and 522, and an inverter 525. The source of the P-type MOS transistor521 is connected to a power source, its gate is connected to an outputterminal Ep 651 of the reference voltage circuit 600, and its drain isconnected to the source of the P-type MOS transistors 523. Furthermore,the gate of the P-type MOS transistor 523 is connected to the Pu of thedigital phase comparator 30. The source of the N-type MOS transistor 522is connected to ground, its gate is connected to an output terminal En626 of the reference voltage circuit, and its drain is connected to thesource of the N-type MOS transistor 524. Furthermore, the gate of theN-type MOS transistor 524 is connected via the inverter 525 to the Pd ofthe digital phase comparator 30. The drains of the P-type MOS transistor523 and the N-type MOS transistor 524 are connected to one another, andthe charge pump output (current monitor) 526 is connected to the loopfilter 52. By obtaining desired voltages from the reference voltagecircuit 600 at En 626 and Ep 651, the current charge pump circuit 520has the function to charge current to the loop filter 52 when Pu is “L,”and to discharge current when Pd is “L.”

FIG. 7 shows the reference voltage circuit 600 used in FIG. 5. Thereference voltage circuit 600 includes a band-gap generation circuit610, an operational amplifier 620, a P-type MOS transistor 650, anN-type MOS transistor 640, and a capacitor 630. The band-gap generationcircuit 610 includes a P-type MOS transistor 619, resistance elements612, 613 and 614, and diodes 615 and 616. The resistance elements 612and 613 have the same resistance values, which is R Ohm. Furthermore,the resistance element 614 has a resistance of r Ohm. The diode 616includes n diodes connected in parallel, each of those diode beingsimilar to the diode 615.

The operational amplifier 620 includes P-type MOS transistors 625, 624and 623, and N-type MOS transistors 621 and 622. The reference voltagecircuit 600 is a negative feedback circuit. With the operationalamplifier 620, the reference voltage circuit 600 compares the voltagesat the nodes 617 and 618, and adjusts the current flowing through theP-type MOS transistor 619 such that they attain the same potential. Thatis to say, when V2 is the voltage at 617, 12 is the current through 613,V1 is the voltage at 618, and I1 is the current through 612, then thefollowing equations are given:V1=V2  (1)I1·R=I2·R  (2)I1=I2  (3)I1=Is=(exp(V1/(n·Vt))−1)  (4)Herein:Vt=kT/q  (5)I2=12·Is(exp(Vd/(n·Vt))−1)  (6)wherein q is the electron charge, k is the Boltzmann constant, and T isabsolute temperature. When Vd is the voltage at the point where theresistor 614 and the diodes 616 are connected, thenV1=r·I2+Vd  (7)n·Vt·log(I1/Is+1)=R·I1+n·Vt·log(I1/(12·Is)+1)  (8)It follows from I1/Is>>1 thatn·Vt·(log(I1/Is)−log(I1/(12·Is)))=R·I1  (9)(n·Vt·log12)/R=I1  (10)That is to say, I1 is proportional to kT/q, and inversely proportionalto the temperature characteristics of R. The capacitor 630 is for phasecompensation of the negative feedback of the reference voltage circuit600.

FIG. 8 shows a configuration example of the switch circuit 930 in FIG.5.

FIG. 9 is a timing chart illustrating the operation of FIG. 6 and FIG.7. In FIG. 9, the horizontal axis denotes time, and the vertical axisdenotes the voltage values of the feedback control signal 3, the inputswitching signal 540, Fp and Fr of the digital phase comparator, and thecharge pump output 526. Before the operation of the PLL circuit 500, thefeedback control signal 3 is “L,” shutting off the feedback loop. Then,by setting the input switch control signal 540 to “L,” the referenceclock 100 is input into Fp of the digital phase comparator 30, and Fr ispegged to “L.” The output voltage of the current charge pump circuit 520rises up to the third clock period, and constantly supplies a current.By monitoring this current or voltage, it can be detected whether thephase comparator 51 and the reference voltage circuit 600 operatenormally.

More specifically, if the capacitor 630 of the reference voltage circuit600 does not have the proper capacitance but has been fabricated smallerthan intended, so that there is no phase margin in the loop of thefeedback system of the reference voltage circuit 600, and this referencevoltage circuit 600 oscillates, then the voltages at En 626 and Ep 651ordinarily have a certain amplitude. In this situation, the currentcharge pump circuit 520 supplies a current corresponding to the voltageamplitude. When the voltage of the charge pump output 526 is monitoredin this case, then ripples occur. These ripples are detected by theripple detection circuit 900, the incremental counter is incremented,and the reference voltage circuit 600 performs a stabilizing operationby increasing the capacitance such that ripples do not occur, In thisexample, it was assumed that the capacitor 630 does not have theappropriate value, but if the reference voltage circuit 600 oscillates,it is also possible to achieve a stable operation from the oscillationwhen the capacitor 630 has the appropriate value with theabove-described configuration.

Embodiment 3

FIG. 10 is an example of a semiconductor integrated circuit according tothe present invention. The PLL circuit 800 in FIG. 10 is almost the sameas the one in FIG. 6, and differs only with regard to the current chargepump circuit 801. The current charge pump circuit 801 in FIG. 10 isalmost the same as the current charge pump circuit 520 in FIG. 6, butthe drains of P-type MOS transistors 806 and 805 are connected to thepoint 804 connecting the P-type MOS transistors 807 and 802. The gatelengths of the P-type MOS transistors 806 and 805 are respectively twotimes and four times that of the P-type MOS transistor 807, and thevarious gates are controlled by bit signals 808 and 809 of a 2-bitregister circuit output bus 840. The gates are connected to switchcircuits 820 that are connected to Ep 651 when those bit signals are“H,” and to a power source when the bit signals are “L.” Furthermore,the drains of N-type MOS transistors 813 and 814 are connected to thepoint 810 connecting the N-type MOS transistors 803 and 812. The gatelengths of the N-type MOS transistors 813 and 814 are respectively twotimes and four times that of the N-type MOS transistor 812, and thevarious gates are controlled by bit signals 815 and 816 of a 2-bitregister circuit output bus 850. The gates are connected to switchcircuits 830 that are connected to En 626 when those bit signals are“H,” and to ground when the bit signals are “L.” The various bits on theregister circuit output buses 840 and 850 are generated from a chargepump output 811 by a voltage differentiating circuit 860, operationalamplifiers 861 and 863, and incremental counters 862 and 864. Vref1 isan upper limiting voltage, and Vref2 is a lower limiting voltage. Itshould be noted that it is also possible to carry out the generation ofthe bits for the respective register circuit output buses 840 and 850from the charge pump output 811 with a tester provided outside thesemiconductor integrated circuit.

FIG. 11 shows a configuration example of the switch circuits 820 in FIG.11, and FIG. 12 shows a configuration example of the switch circuits 830in FIG. 10.

FIG. 13 is a timing chart illustrating the operation of FIG. 10. In FIG.13, the horizontal axis denotes time, and the vertical axis denotes thevoltage values of the feedback control signal 3, the input switchingsignal 540, Fp and Fr of the digital phase comparator 30, and the chargepump output 511, as well as the current value of the charge pump 811.FIG. 13 illustrates the case that the characteristics of the P-type MOStransistor 807, which serves as the current source for the currentcharge pump circuit, are poor. Before the operation of the PLL circuit,the feedback control signal 3 is “L,” shutting off the feedback loop.Then, by setting the input switch control signal 540 to “L,” thereference clock 100 is input into Fp of the digital phase comparator 30,and Fr is pegged to “L.” The voltage 811 of the current charge pumpcircuit 801 rises up to the third clock period, and constantly suppliesa current. However, in the first period, the current value of thecurrent charge pump circuit 801 is smaller than the appropriate currentvalue. Thus, the register output 840 is shifted, and by setting “00” to“01”, the current value of the current charge pump circuit 801 assumesthe appropriate value in the second period. Furthermore, by setting theinput switch control signal 540 to “H” in the fourth period, thereference clock signal 100 is input into Fr of the digital phasecomparator 30, and Fp is pegged to “L.” The voltage 811 of the currentcharge pump circuit 801 decreases, and the current is constantlydischarged. Since the current value is already appropriate in the fourthperiod, the register output 850 is sustained at “00”. Thus, it ispossible to attain an appropriate current value by monitoring thischarge pump circuit, and adjusting the current source of the charge pumpcircuit with the incremental counters 862 and 864, and thus it becomespossible to reduce tiny current variations, such as those caused byprocess variations. It should be noted that this example has beenexplained only for a P-type MOS transistor, but a similar approach isalso suitable for deterioration of N-type MOS transistors, that is, fordischarge.

Embodiment 4

FIG. 14 illustrates another example of a semiconductor integratedcircuit in accordance with the present invention. Numeral 400 denotes asemiconductor integrated circuit in accordance with the presentinvention. Numeral 480 denotes a clock distribution circuit connected toa switch circuit 420. In response to a bypass control signal 473, theswitch circuit 420 switches between the reference clock 100 that isinput into a PLL circuit 50 and the clock that has been multiplied witha PLL circuit 50. The clock distribution circuit 480 distributes clocksover the clock lines 430, 431 and 432 to function blocks A, B and C.Respective drivers 485 a and 485 b of the clock lines 431 and 432 havethe function to increase or decrease the driver intensity withcorresponding output buses 441, 442, 443 and 444 of control registercircuits 490. The respective clock lines 430, 431 and 432 are connectedto phase detectors 410 detecting rising edges, one of which is a phasedetector 460 detecting phase differences between the clock lines 430 and431, and supplying an up signal 461 and a down signal 462 to one controlregister circuit 440. The other one is a phase detector 470 detectingphase differences between the clock lines 431 and 432, and supplying anup signal 471 and a down signal 472 to the other control registercircuit 450. Numeral 463 denotes a comparison termination signal that isapplied by the one control register circuit 440 to the other controlregister circuit 450.

FIG. 15 is an example of the phase comparator 410, which is made ofinput ports Fp and Fr, inverters 411 and 412, 2-input NAND circuits 413,414, 415 and 416, and output ports Up and Dn. The reference clock isinput from Fp, and is input into the inverter 411 and the NAND circuit413. Furthermore, also the output from the inverter 411 is input intothe NAND circuit 413. The clock to be compared is input from Fr, and isinput into the inverter 412 and the NAND circuit 414. Furthermore, alsothe output from the inverter 412 is input into the NAND circuit 414. The2-input NAND circuits 415 and 416 constitute an R-S latch circuit, whichdetects falling edges in the output of the NAND circuits 413 and 414. Ifthe rising edge of Fr lags behind the rising edge of Fp, then the Upoutput becomes “H,” for the time of that phase difference delay. If therising edge of Fr leads the rising edge of Fp, then the Dn outputbecomes “L” for the time of that phase difference delay.

FIG. 16 is an example of the switch circuit 420, which is made of acontrol signal port e, two input ports i1 and i2, an output port o, aninverter 424, P-type MOS transistors 425 and 428, and N-type MOStransistors 426 and 427. When the input port e is “H,” then i2 is outputat output port o, and when the input port e is “L,” then i1 is output atoutput port o.

FIG. 17 shows an example of the control register circuit 490. Thecontrol register circuit 490 is made of a comparison terminationdetection circuit 300, incremental registers 493 and 494, input ports R,CK, Up, Dn, and output ports Eo, Uo and Do. The input port R resetsignal 492 is connected to the input ports R of the comparisontermination detection circuit 300 and the incremental registers 493 and494. The input port CK is input into the comparison terminationdetection circuit 300, and the input port Up is input via a dynamiccircuit 499 into an input port in of the incremental register 493 and aninput Din of the comparison termination detection circuit 300. Inputport Dn is connected via an inverter 487 and a dynamic circuit 488 toinput port in of the incremental register 494 and input port Din2 of thecomparison termination detection circuit 300. In the dynamic circuit488, numeral 485 denotes an N-type MOS transistor, and numeral 486denotes a P-type MOS transistor. Output port Eo is connected to out1 ofthe comparison termination detection circuit 300, output port Uo isconnected to output ports O1 and O2 of the incremental register 493, andoutput port Do is connected to output ports O1 and O2 of the incrementalregister 494. The incremental registers 493 and 494 are made of a serialconnection of 1-bit incremental registers 496, which include a HA and aflip-flop with reset. The 1-bit incremental registers 496 have inputports in, CK and R, and output ports O2 and O1. A clock 491 is inputinto CK, and the reset signal 492 is input into R. The output port O1 isthe output of the flip-flop, whereas O2 is a carry signal.

The comparison termination detection circuit 300 is very similar to theripple elimination termination circuit 220 of Embodiment 1, and FIG. 18shows an example. The comparison termination detection circuit 300 inFIG. 18 includes flip-flops 303, 304, 305 and 360 with reset, EXORcircuits 312 and 313, a 4-input AND circuit 311, AND circuits 314 and318, an OR circuit 315 and an inverter 317. When the states of the Upsignal and the Dn signal, which are the signals input into the controlregister circuits 490 do not change within two periods of the referenceclocks, or when the Up signal and the Dn signal have changed todifferent states within three periods, then a comparison terminationsignal (Eo) is output from out1, the clocks (clocka and clockb) 489 and491 that are used within the control register circuits 490 are stopped,and the respective contents of the incremental registers 493 and 494 arehold.

FIG. 19 is a timing chart illustrating FIGS. 14, 15 and 17. In FIG. 19,the horizontal axis denotes time, and the vertical axis denotes thevoltage values of various signals, namely the bypass control signal 473,the reference clock 100, the clock signal line 430 supplied to functionblock A, the clock signal line 431 supplied to function block B, theclock signal line 432 supplied to function block C, the output ports Upand Dn of the phase comparator 460, the output ports Up and Dn of thephase comparator 470, the output bus of the control register circuit440, and the output bus of the control register circuit 450. In thisexample, the rising edge of the clock signal line 431 supplied to thefunction block B lags behind the rising edge of the clock signal line430 supplied to the function block A, and the rising edge of the clocksignal line 432 supplied to the function block C lags behind the risingedge of the clock signal line 431 supplied to the function block B.First, when the PLL circuit starts its stabilizing operation, the PLLONsignal 56 is turned from “L” to “H,” and the reset signals of thecontrol register circuits 440 and 450 are released. The bypass controlsignal 473 is “L,” and the PLL circuit 50 performs an internal feedbackloop control and starts preparations for the stabilization operation.

The reference clock 100 is supplied to the clock distribution circuit480, and the clock phase difference between the clock signal lines 430and 431 is detected by the phase detector 460. During the first period,the rising edge of the clock on 431 lags behind that of 430, so that theUp output of the phase comparator 460 becomes “H.” Thus, the first bitUo[0] of the incremental register 493 of the control register circuit440 becomes “H,” strengthening the driver 485 a for the clock line 431.During the second period, there is no phase difference between the clocklines 430 and 431, and the Up output of the phase comparator 460 stays“L,” and the Dn output stays “H.” Also in the third period, there is nophase difference between the clock lines 430 and 431, so that clockdistribution without phase difference is possible. Then, the controlregister circuit 440 outputs the comparison termination signal 463, andthe reset of the control register circuit 450 is released. Next, thephase comparator 470 starts to compare the phase difference between theclock lines 432 and 431. During the fourth period, the Up output of thephase comparator 470 becomes “H.” Thus, the first bit Uo[0] of theincremental register 493 of the control register circuit 450 becomes“H,” strengthening the driver 485 b for the clock line 432. During thefifth period, the Dn output of the phase comparator 470 is “L,” and thefirst bit Do[0] of the incremental register 494 of the control registeroutput 450 becomes “H” (not shown in the drawings), reducing thecapability of the driver 485 b of the clock line 432. During the sixthperiod, the Up output of the phase comparator 470 again becomes “H.” Thephase difference between the clock lines 432 and 431 cannot be made anysmaller than that, so that the control register circuit 450 outputs thecomparison termination signal 463, the bypass control signal 473 becomes“H” in the seventh period, and the output signal of the PLL circuit 50is supplied from the clock distribution circuit 480 to the functionblocks.

Thus, before the PLL circuit 50 starts its stabilizing operation, clockskewing of the function blocks can be eliminated by adjusting thestrength of the clock drivers 485 a and 485 b of the clock distributioncircuit 480, so that it becomes possible to adjust the clock phases ofthe semiconductor integrated circuit 400 with high precision.

Embodiment 5

FIG. 20 is an example of another semiconductor integrated circuitaccording to the present invention, which includes a PLL circuit 50operated with a reference clock 100, a clock supply circuit 60, a switchcircuit 420, and an SRAM (static random access memory) circuit 700. Theclock supply circuit 60 is connected to the output of the PLL circuit50. Using a bypass control signal 703, the switch circuit 420 switchesbetween the reference clock 100 and the output of the clock supplycircuit 60. The SRAM circuit 700 is synchronized with the output of theswitch circuit 420. The SRAM circuit 700 has an address 741 as an inputport, and an SRAM data output 763 and a bypass control signal 703 asoutput ports. Furthermore, he SRAM circuit 700 includes an addressdriving circuit 740, a memory access circuit 710, a precharge array, asense amplifier array 760, a comparator 770, an incremental/decrementalregister 750, and a sense amplifier activation signal generation circuit780. The address driving circuit 740 drives an address signal line 742in correspondence with an address 741. The memory access circuit 710 ismade of a memory cell array 730 and a row decoder array 720. Theprecharge array precharges a bit line pair 711 of the memory cell array730. the sense amplifier array 760 amplifies the voltage of the bit linepair 711. The comparator 770 compares the output 761 of the senseamplifier array 760 with a reference voltage. Theincremental/decremental register 750 stores the state of the output 771of the comparator 770 in synchronization with the reference clock 100.The sense amplifier activation signal generation circuit 780 controlsthe delay time of an activation signal 781 for the sense amplifier array760 with the output state of the incremental/decremental register 750.The output of the switch circuit 420 is given via a buffer 701 and abuffer output signal line 702 to the memory access circuit 710, and viathe sense amplifier activation signal generation circuit 780 to thesense amplifier array 760. Numerals 782, 783, 784 and 785 are delaycircuits (inverters) in the sense amplifier activation signal generationcircuit 780. The output 762 of the sense amplifier array 760 passesthrough an output circuit array before becoming the SRAM data output763.

FIG. 21 is an example of the memory access circuit 710. The memoryaccess circuit 710 includes a dummy memory cell array having N columnsof dummy memory cells 731, a row decoder 721 (see FIG. 22), a memorycell array 730 made of N columns×M rows of memory cells 732, and M rowdecoders 722 (see FIG. 23). The row decoder 721 constantly activates thedummy word line 723 in synchronization with the clock when the bypasscontrol signal 703 is inactivated. The row decoders 722 activate therespective word lines 724 with the status of the address 741 insynchronization with the clock when the bypass control signal 703 isactivated. In FIGS. 22 and 23, numeral 725 denotes an AND circuit,numeral 726 denotes a decoding circuit, and numeral 727 denotes aninverter.

The dummy memory cells 731 are circuits as shown in FIG. 24, and havethe function to transmit the bit information “0” within the memory cellto a bit line pair (BL, BLB) 712 when the word line (WD) 723 isactivated.

The regular memory cells 732 are circuits as shown in FIG. 25, and havethe function to transmit the bit information within the memory cell tothe bit line pair (BL, BLB) 712 when the word line (WD) 724 isactivated.

FIG. 26 shows a sense amplifying circuit 764 that constitutes one bitportion of the sense amplifying array 760. The sense amplifying circuit764 in FIG. 26 includes N-type MOS transistors 746, 747 and 779, as wellas P-type MOS transistors 765, 766, 777 and 778, and a sense amplifieroutput line 749.

FIG. 27 is an example of the comparator 770. The EXOR circuits 772, 773and 774 compare the ground signal (expectation value) with the output oof the sense amplifying circuits 764 in the sense amplifying array 760connected to the first column, the N/2-th column and the N-th column ofthe dummy memory cell array. The output of the EXOR circuits 772, 773and 774 is input into a 3-input AND circuit 775, and a comparison outputsignal 771 is obtained from a latch 219 that operates in synchronizationwith a clock 758.

FIG. 28 is an example of the incremental/decremental register 750. Theincremental/decremental register 750 is made of an inverter 741, a phasecomparison termination circuit 200, a 2-bit incremental/decrementalregister 743, input ports R, CK, Up, and output ports Eo and Uo. A resetsignal 759 for the input port R is also connected to the input ports Rof the phase comparison termination circuit 200 and theincremental/decremental register 743. The input port CK is input intothe phase comparison termination circuit 200. And the input port Up,which receives the comparison output signal 771 is input into the port“in” of the incremental/decremental register 743, and the port Din ofthe phase comparison termination circuit 200. The output clock (clockb)758 of the phase comparison termination circuit 200 is connected to theclock input port of the incremental/decremental register 743. A 1-bitlogic circuit 753 is constituted by AND circuits 756, 754, and aninverter 742. Numeral 752 denotes 1-bit incremental/decremental registercircuits, which are made of a 1-bit logic circuit 753 and a flip-flop757 with reset. Numeral 743 denotes the 2-bit incremental/decrementalregister circuit, including two 1-bit incremental/decremental registercircuits 752 connected in series. The output bus Uo 751 is made of theinverted lower bit and the upper bit.

FIG. 29 is a timing chart explaining FIG. 20. In FIG. 29, the horizontalaxis denotes time, and the vertical axis denotes the voltage values ofvarious signals, namely the bypass control signal 703, the referenceclock 100, the dummy word line 723, the bit line pair 711, thecomparator output 771, the sense amplifier activation signal 781, andthe output bus 751 of the incremental/decremental register 750. When thesignal starting the operation of the PLL circuit 50, that is the PLLONsignal 56 becomes “H,” the reset of the flip-flop 757 within theincremental/decremental register 750 is released. At first, the bypasscontrol signal 703 is “L,” so that the reference clock 100 is connecteddirectly to the SRAM circuit 700. Then, the dummy word line 723 rises,and the internal bit information “0” of the dummy memory cells 731 istransmitted to the bit line pairs 712 of the dummy memory cells 731, adifference occurs in the voltage of the bit line pairs 711, and thesense amplifier activation signal 781 is activated. The comparator 770performs a comparison thereof, and in this example, since the comparisonresult of the first period is that they are different, there is anincrement, and the output bus 751 of the incremental/decrementalregister 750 outputs “01.” Thus, the delay of the driver of the senseamplification signal 781 is increased, and regular operation becomespossible at the second period.

Regular operation is also performed at the third period, and the bypasscontrol signal 703 from the phase comparison termination circuit 200becomes “H.” Furthermore, the internal content of theincremental/decremental register 750 is held, and the clock from theclock supply circuit 60 is supplied to the SRAM circuit 700.

As described above, racing errors of the sense amplification signal 781and the word line can be eliminated before the PLL circuit 50 reachesstable operation, and it becomes possible to attain a highly preciseSRAM circuit 700 and semiconductor integrated circuit.

Embodiment 6

FIG. 30 is an example of a semiconductor integrated circuit according toanother embodiment of the present invention. A data holding circuit 70in FIG. 30 includes a circuit 81 in which two stages of dynamic circuits92 and 93 are connected in series, and a switch circuit 420 thatswitches between the reference clock 100 and the output of a clocksupply circuit 60, depending on a bypass control circuit 90. The firstdynamic circuit 92 is made of N-type MOS transistors 71, 72, 73 and 74,and a P-type MOS transistors 75, and receives a clock 85 from the switchcircuit 420. When the bypass control signal 90 is inactivated, theN-type MOS transistor 74 in the first dynamic circuit 92 is turned onand off in synchronization with the clock 85, and the N-type MOStransistors 71, 72 and 73 are constantly off. When the bypass controlcircuit 90 is activated, the gates of the N-type MOS transistors 71, 72and 73 are connected to ordinary data lines 87, 88 and 89. Thesecond-stage dynamic circuit 93, which is connected to an output node 94of the first dynamic circuit 92, is made of N-type MOS transistors 77and 78, a P-type MOS transistor 76, and an inverter 79, and receives aclock 91 from a delay adjustment circuit 84. A comparator 80 comparesthe output 82 of the second-stage dynamic circuit 93 with an expectationvalue, and a comparator output 83 that is held by a latch 219 operatingin synchronization with the clock 758 is supplied to the controlregister (incremental/decremental register) 750. Then, the strength ofthe driver in the delay adjustment circuit 84 for the clock 91 that isgiven into the second-stage dynamic circuit 93 can be increased with theoutput bus 86 of that control register 750.

FIG. 31 is a timing chart explaining FIG. 30. In FIG. 31, the horizontalaxis denotes time, and the vertical axis denotes the voltage values ofvarious signals, namely the bypass control signal 90, the referenceclock 100, the clock signal 85 of the first-stage dynamic circuit, theclock signal 91 of the second-stage dynamic circuit, the dynamic circuitoutput signal 82, the output signal 83 of the comparator, and the outputsignal 86 of the incremental/decremental register 750. When the PLLONsignal 56 becomes “H,” the reset of the incremental/decremental register750 is released. Then, since the bypass control signal 90 is “L,” theclock 85 of the first-stage dynamic circuit is connected directly to thereference clock 100. Moreover, since the bypass control signal 90 is“L,” the N-type MOS transistor 74 is turned on and off, and the N-typeMOS transistors 71, 72 and 73 are off. At the clock of the first period,the dynamic circuit output 82 becomes “H.” Up to now, it should be “L.”The comparator circuit 80 outputs “H,” and the register output 86 ischanged from “01” to “10.” Thus, the delay of the clock 91 of the seconddynamic circuit is increased. In the second period, the dynamic circuitoutput 82 becomes “L,” and ordinary operation becomes possible. Then inthe third period, there is another miss, and in the fourth period, thereis a hit. Then, the incremental/decremental register 750 sets the bypasscontrol signal 90 to “H” and holds the register internal information,and the dynamic circuit 81 is directly connected to the output of theclock supply circuit 60.

As described above, the delay of the clock 91 is adjusted such that thesecond-stage dynamic circuit 93 is activated after the potential of theoutput node 94 of the first-stage dynamic circuit 92 has settled. Thus,racing errors in the two-phase clock of the serially connected dynamiccircuit 81 can be eliminated until the PLL circuit has stabilized, andit is possible to realize a highly accurate semiconductor integratedcircuit.

In the above-described Embodiments 4 to 6, if another type of clockgeneration circuit is used instead of the PLL circuit 50, the adjustmentof the portion corresponding to those in the above embodiments iscarried out using the reference signal 100 before that clock generationcircuit supplies a system clock signal.

Embodiment 7

FIG. 32 is another example of a semiconductor integrated circuit inaccordance with the present invention. Numeral 1000 denotes asemiconductor integrated circuit. Numeral 1010 denotes a cache circuitthat is synchronized with a clock when a block reset signal is released,and that includes a tag portion 1020 and a data portion 1040. The tagportion 1020 is made of an SRAM circuit 1025 and a comparator circuit1030. The tag portion 1020 reads an upper address from the SRAM circuit1025 storing the upper address within the tag at a lower address, andcompares the upper address coming from an external block with thecomparator circuit 1030. A data portion 1040 accesses an internal memoryat the lower address, receives a hit signal 1031 from the tag portion1020, and has the function to output or write data when the hit signal1031 indicates a hit. Furthermore, the data portion 1040 includes asense amplifier and an output circuit, and has a circuit 1041 thatcontrols with a register signal 1052 whether an activation signal 1043for the sense amplifier and an output activation signal 1044 shouldoperate in response to the hit signal 1031, or whether they shouldoperate in synchronization with the regular clock. Moreover, thesemiconductor integrated circuit 1000 also includes a function block Cthat is synchronized with a clock 61 when the block reset signal isreleased, and which includes a comparator circuit 1060 that takes in theoutput data from the data portion 1040 of the internal cache at theclock 61, and compares it with an expectation value. The comparatorcircuit 1060 is also provided with the function to hold the internalcontent of the first clock period. The control register 1050 issynchronized with the clock 62, the reset of its internal registers isreleased with a phase fine-tuning period transmission signal 1071, andits internal registers are incremental counters. When the output signal1061 of the comparator circuit 1060 is “L,” then it operates insynchronization with the clock, and when it is “H,” its operation stops.Moreover, it outputs a stop signal (Eo) 1051.

The semiconductor integrated circuit 1000 further has a phasefine-tuning period transmission circuit 1070, which has the function totransmit the fact that the phase fine-tuning period has been entered atthe time when the phase fine-tuning period has been entered after thecapturing period of the PLL 50. FIG. 33 is an example of the phasefine-tuning period transmission circuit 1070, which is made of a 1:4frequency divider 1072 in synchronization with the reference clock, afour-bit incremental register and OR circuit 1073, and a flip-flop 1074.When any of the upper two bits of the incremental register circuit 1073is “H,” then an “H” is given out as 1071, thereby transmitting the factthat a phase fine-tuning adjustment period has been entered. It shouldbe noted that the incremental registers 259 of the various bitsconstituting the incremental register circuit 1073 have the sameinternal configuration as shown in FIG. 3.

The phase fine-tuning period transmission signal 1071 releases the resetof the control register 1050 inside the data portion 1040. Furthermore,only when the block reset signal is “L” and the phase fine-tuning periodtransmission signal 1071 is “H,” the cache circuit 1010 accesses thedummy memory cells, and the comparator circuit 1030 hits at every cycle,and access and read-out of the dummy memory cells are performed at eachcycle in the data portion 1040. The dummy memory cells are circuitshaving the same function as those in FIG. 24 described above.

FIG. 34 shows a configuration example of the switch circuit 1042 in FIG.32.

FIG. 35 is a timing chart explaining FIG. 32. In FIG. 35, the horizontalaxis denotes time, and the vertical axis denotes the voltage values ofvarious signal lines, namely the block reset signal, the phasefine-tuning period transmission signal 1071, the PLL feedback signal Fr,the tag hit signal 1031, the cache data portion dummy word line 723, thesense amplifier activation signal 1043, the output activation signal1044, and the comparator circuit output signal 1061. When the PLLcircuit 50 is started up, the block reset signal is “L” until it reachesstable oscillation, and any data access to the function blocks isinvalid. When the PLL circuit 50 enters the phase fine-tuning period,the phase fine-tuning period transmission signal 1071 becomes “H,” andis supplied to the cache circuit 1010.

The comparator circuit 1060 is always synchronized with the clock 61 andoutputs a clock that is delayed by the memory access time of the tagportion 1020.

The dummy word line 723 of the data portion 1040 operates ordinarilyonly during the phase fine-tuning period. As for the register output1052 of the first period, the sense amplifier activation signal 1043acts depending on the tag bit signal 1031, and the output activationsignal 1044 acts only in synchronization with the clock 62. In thisexample the comparator circuit 1060 misses, and detects that it isimpossible to output normal data with the sense amplifier activationsignal 1043 generated by the tag hit signal 1031. Then in the secondperiod, the output of the control register 1050 is changed from “01” to“10.”

The sense amplifier activation signal 1043 is synchronized with theclock 61, and operates after the output activation signal 1044 hasreceived the tag hit signal 1031. However, in the third period, thecomparator circuit 1060 hits, and this time detects that it is possibleto output normal data with the output activation signal 1044 generatedby the tag hit signal 1031. Then, the control register 1050 holds itscontent.

As described above, when the tag portion indicates a cache miss duringordinary operation, whether the operation of either the sense amplifyingcircuit or the output circuit within the data portion 1040 should bestopped is determined in accordance with the oscillation frequency ofthe PLL circuit 50 at a time when that frequency has settled. Morespecifically, if the clock frequency is low, then the operation of thesense amplifier is stopped, and if it is high, then operation of thesense amplifier is allowed but the operation of the output circuit isstopped. Thus, in accordance with clock frequency, device conditions,and temperature dependency, it is possible to stop the optimal logicportion when stopping invalid data during one cycle of the clock, inorder to reduce the power consumption. This means, a semiconductorintegrated circuit whose power consumption can be reduced efficientlycan be achieved.

It should be noted that in the above-described embodiments, thereference clock signal 100 may be supplied from an internal oscillationcircuit within the semiconductor integrated circuit, or it may besupplied from outside the semiconductor integrated circuit.

The invention may be embodied in other forms without departing from thespirit or essential characteristics thereof. The embodiments disclosedin this application are to be considered in all respects as illustrativeand not limiting. The scope of the invention is indicated by theappended claims rather than by the foregoing description, and allchanges which come within the meaning and range of equivalency of theclaims are intended to be embraced therein.

1-14. (canceled)
 15. A semiconductor integrated circuit comprising; aclock generation circuit configured for generating a system clock signalfrom a reference clock signal; a switch circuit configured forselectively outputting the reference clock signal or the system clocksignal; and a clock distribution circuit configured for feeding a clocksignal outputted from the switch circuit to a plurality of circuitslocated outside of the clock generation circuit; wherein the switchcircuit outputs the reference clock signal until the clock generationcircuit starts stabling operation, and the switch circuit detects thatthe clock generation circuit starts stabling operation by a controlsignal inputted into the switch circuit.
 16. The semiconductorintegrated circuit according to claim 15, wherein the control signal isgenerated based on comparison of phases of a plurality of clock signalsoutputted from the switch circuit.
 17. The semiconductor integratedcircuit according to claim 16, wherein the control signal is generatedbased on a comparison termination signal indicating comparison result ofphases of a plurality of clock signals outputted from the switchcircuit.
 18. The semiconductor integrated circuit according to claim 17,wherein the switch circuit outputs the system clock signal instead ofthe reference clock signal after the comparison termination signalindicates that difference of phase between the plurality of clocksignals outputted from the switch circuit cannot be made any smaller.